Prior art receivers for fiber optic signals have usually used fixed gain and alternating current coupling techniques. Any attempts automatically to control the gain of such receivers have been slow in response, and therefore inadequate. Also, the systems used in such prior art receivers for tracking signal level in the presence of large direct current drifts and offsets have used alternating current coupled amplifier stages, which have not proven to be satisfactory.
Drifts in the electrical signals in the receiver derived from converting the received optical signals which results, for example, from photodiode dark current, or amplifier offset voltages, can exceed the lowest signal levels in the receiver dynamic range. Signal base line stability must be maintained despite these drifts, because the final signal detection is essentially an amplitude identification process. The prior art receivers, however, are not capable of maintaining signal base line stability. When direct-coupled amplifiers are used in the prior art receivers, drift currents exceed the lower signal levels. On the other hand, alternating current coupled amplifiers gradually shift the signal base line to the half amplitude signal direct current component. Direct current coupled amplifiers with final alternating current coupled and direct current restoration output stages appear to be feasible. However, all these systems are susceptible to the effects of the receiver adjustable gain stage, which is required for the wide signal range to be processed by the receiver. The apparent voltage offset changes produced by the adjustable gain stage have a tendency to cause unwanted base line modulations.
In general, the large amplitude range that must be handled by receivers for unipolar trapezoidal Manchester fiber optic signals complicates the signal detection process. The two levels of the unipolar trapezoidal signal waveform and, especially the timing of the transitions between these levels, contain the essential data information. The large amplitude signals overdrive the amplifiers and limiters in the receivers, and this produces false width and transition distortions which can cause decoding errors.
As noted, the unipolar trapezoidal Manchester fiber optics signal is symmetrical about its half amplitude level. The comparator in the receiver detector provides an ideal detection stage if it switches the receiver output when the Manchester signal crosses the half peak amplitude level which is the best decision threshold. Signal amplitude at the comparator varies, however, in a transient manner during message start-up, and over steady stage line levels for wide amplitude ranges. A direct current threshold, or a threshold based merely on signal history, would not then occur at the half amplitude for every decision. This disparity between the signal half amplitude and the threshold causes additional errors in the prior art receivers.
Although electrical multi-terminal systems transmit a fairly constant, high amplitude, bipolar electrical signal to the various receivers, the fiber optics transmission system is unipolar in that information is represented by the presence or absence of optical data. Signal amplitude can vary over wide ranges because of system loss variations. The front end signal-to-noise ratio of the receiver defines the lower limit of the dynamic range of the signal. The unipolar Manchester optical signal has a trapezoidal shape with respect to the zero base line and therefore, a threshold set at half the signal peak amplitude is required in order to reproduce accurately the pulse width, this being essential in order to decode the signals.
Therefore, in the construction of a receiver suitable for the detection of trapezoidal unipolar Manchester fiber optic signals, certain critical aspects of the signals must be considered. These include the fact that the manchester optical signal is a unipolar signal with zero base line; that it has a trapezoidal wave shape which is symmetrical around the half peak amplitude level; that it is a low amplitude signal with high dynamic range (approximately 100:1); and that it has low transmission noise and wide transmission bandwidth.
Detection algorithms in the receiver are susceptible to noise at signal transitions as well as at peak amplitudes, and they rely on pulse-width accuracies for acceptable bit error rate performance. Accordingly, the receiver must exhibit low front end noise, good base line stability, linear or predictable amplitude response over a wide range, and an accurate decision threshold at the half peak amplitude signal levels.
Because of fiber optics inherent immunity to electromagnetic interference, the dominant noise sources in the fiber optic communication system occur at the input to the receiver. The receiver input includes a photo-detector stage, and noise currents are related to the reverse currents of the photodiode in that stage, and to the reverse currents of the gate of the field effect transistor also included in the photo-detector stage, by the expression: EQU i.sub.N.sup.2 =2qi.sub.r .DELTA.f (1)
where:
q=the electronic charge, 1.6019.times.10.sup.-19 coulomb PA1 i.sub.r =reverse current PA1 .DELTA. f=noise bandwidth, Hz PA1 K=Boltzman's constant, 1.38.times.10.sup.23 joules/degree Kelvin PA1 T=temperature, degrees Kelvin PA1 R=resistance or, for FET's 1/gm=R PA1 .DELTA. f=bandwidth, Hz
Amplifier input noise and resistor noise voltages are similar in form: EQU E.sub.n.sup.2 =4KTR .DELTA.f (2)
where:
Because the optical input signal is an on-off, or unipolar signal, a linear mix of the signal and noise implies that the signal should be threshold detected at half peak amplitude for balanced exclusion of noise peaks.
The receiver of the invention makes a significant contribution to Manchester optical signal decoding in the presence of noise. A reduction in receiver bandwidth to about 1.25 times the bit ratio is effective to improve the signal-to-noise ratio and consequently reduces the bit error rate. Linear phase filters are used in the receiver, and these filters exhibit excellent phase and amplitude response, and have a linear phase characteristic which does not add any distortion to random phases or Gaussian noise. The filters also decrease the peaks of impulse noise to those of the signal frequencies within the filter pass band.
In the Manchester code, a 1,0,1 bit pattern has twice the period of either the 1,1,1, or 0,0,0 bit patterns. An optimum noise bandwidth filter is included in the receiver of the invention to convert the higher frequency square waves of the Manchester signal into sine waves; while the alternating patterns retain their trapezoidal shapes. Therefore, after propagation through low pass filters, the dissimilar bit sequences have different instantaneous signal-to-noise ratios. The receiver of the invention also includes means for preserving or predicting the half peak amplitude level of the Manchester signals so as to obviate amplitude width or transition distortion.
To reiterate, the zero crossings of the bipolar signals are equivalent to the half peak amplitude crossings of the Manchester signals. For the Manchester signals, errors in base line reference, thresholds, or signal half peak amplitude levels produce output widths or transition errors. The receiver of the invention is constructed to eliminate such errors. Stability of the signal base line provides the threshold and comparison circuits of the receiver with a reference to measure the input signal at the half peak amplitude level. Some of the significant sources of errors affecting the base line stability are photodiode dark current, amplifier offsets, drifts and leakages, and signal offsets. The receiver of the invention is constructed to compensate for errors produced by the foregoing error sources.